Apparatus and method for multilevel coding (mlc) with binary alphabet polar codes

ABSTRACT

A method includes receiving multiple bits to be transmitted. The method also includes applying a first binary alphabet polar code to a first subset of the multiple bits to generate first encoded bits. The first encoded bits are associated with a first bit level of a multilevel coding scheme. The method further includes generating one or more symbols using the first encoded bits and bits associated with a second bit level of the multilevel coding scheme. The first binary alphabet polar code is associated with a first coding rate. In addition, the method could include applying a second binary alphabet polar code to a second subset of the multiple bits to generate second encoded bits. The second encoded bits are associated with the second bit level. The second binary alphabet polar code is associated with a second coding rate such that the bit levels have substantially equal error rates.

CROSS-REFERENCE TO RELATED APPLICATION AND PRIORITY CLAIM

This application is a continuation application of U.S. patentapplication Ser. No. 16/152,754, which is a divisional of U.S. patentapplication Ser. No. 14/503,547, filed Oct. 1, 2014, which also claimspriority under 35 U.S.C. § 119(e) to U.S. Provisional Patent ApplicationNo. 61/885,351 filed on Oct. 1, 2013. This provisional patentapplication is hereby incorporated by reference in its entirety.

TECHNICAL FIELD

This disclosure is generally directed to communication systems. Morespecifically, this disclosure is directed to an apparatus and method formultilevel coding (MLC) with binary alphabet polar codes.

BACKGROUND

Many modern wireless and other communication systems encode digital datafor transmission over communication paths. One type of coding mechanisminvolves the use of polar codes, which are used to encode data bits intoencoded data bits that are then mapped into symbols for transmission.Polar codes can be used to achieve capacity of binary-input memorylesssymmetric channels, meaning the polar codes allow the full capacity of achannel to be utilized. Polar codes allow this to be achieved usinglow-complexity encoding and decoding algorithms.

Recently, various attempts have been made to use communication channelsmore efficiently based on higher-order modulation techniques, such asM-ary quadrature amplitude modulation (M-QAM). The phrase “M-ary”modulation generally refers to modulation where there are Mconstellation points so that one symbol carries log₂(M) bits.

Extending the use of polar codes to systems that use higher-ordermodulation techniques is not a simple task. For example, output bitsfrom a binary polar code often experience different effective channelsbecause of their mapping to higher-order symbols. As a result, standardtechniques for polar code design are not generally applicable to systemsusing higher-order modulation techniques.

SUMMARY

This disclosure provides an apparatus and method for multilevel coding(MLC) with binary alphabet polar codes.

In a first embodiment, a method includes receiving multiple bits to betransmitted. The method also includes applying a first binary alphabetpolar code to a first subset of the multiple bits to generate firstencoded bits. The first encoded bits are associated with a first bitlevel of a multilevel coding scheme. The method further includesgenerating one or more symbols using the first encoded bits and bitsassociated with a second bit level of the multilevel coding scheme. Thefirst binary alphabet polar code is associated with a first coding rate.

In a second embodiment, a first encoder is configured to (i) receivemultiple bits to be transmitted and (ii) apply a first binary alphabetpolar code to a first subset of the multiple bits to generate firstencoded bits. The first encoded bits are associated with a first bitlevel of a multilevel coding scheme. The apparatus also includes asymbol generator configured to map the first encoded bits and bitsassociated with a second bit level of the multilevel coding scheme toone or more symbols for transmission. The first binary alphabet polarcode is associated with a first coding rate.

In a third embodiment, a method includes applying a first binaryalphabet polar code to first decoding information associated with one ormore symbols to generate first decoded bits. The first decodinginformation is associated with a first bit level of a multilevel codingscheme. The method also includes outputting the first decoded bits andbits associated with a second bit level of the multilevel coding scheme.The first binary alphabet polar code is associated with a first codingrate.

In a fourth embodiment, an apparatus includes a first decoder configuredto apply a first binary alphabet polar code to first decodinginformation associated with one or more symbols to generate firstdecoded bits. The first decoding information is associated with a firstbit level of a multilevel coding scheme, and the multilevel codingscheme has a second bit level. The first binary alphabet polar code isassociated with a first coding rate.

In a fifth embodiment, a method includes identifying a target error rateassociated with a multilevel coding scheme, where the multilevel codingscheme has multiple bit levels. The method also includes identifying aneffective signal-to-noise ratio (SNR) at each of the bit levels, wherethe effective SNR is based on a symbol mapping rule used to map bitsinto symbols. The method further includes identifying a code rate ateach bit level using the target error rate and the SNR at that bitlevel. In addition, the method includes selecting, at one or more of thebit levels, at least one polar code having at least one of theidentified code rates so that an overall error rate of each bit level issubstantially equal.

Other technical features may be readily apparent to one skilled in theart from the following figures, descriptions, and claims.

BRIEF DESCRIPTION OF THE DRAWINGS

For a more complete understanding of this disclosure and its features,reference is now made to the following description, taken in conjunctionwith the accompanying drawings, in which:

FIGS. 1A and 1B illustrate example systems that use multilevel coding(MLC) with binary alphabet polar codes in accordance with thisdisclosure;

FIGS. 2 through 7 illustrate example encoding and decoding schemessupporting MLC with binary alphabet polar codes in accordance with thisdisclosure;

FIGS. 8A and 8B illustrate example methods for using MLC with binaryalphabet polar codes in accordance with this disclosure;

FIG. 9 illustrates an example method for designing an MLC coding schemewith binary alphabet polar codes in accordance with this disclosure; and

FIG. 10 illustrates an example device supporting the design of an MLCcoding scheme with binary alphabet polar codes in accordance with thisdisclosure.

DETAILED DESCRIPTION

FIGS. 1A through 10, discussed below, and the various embodiments usedto describe the principles of the present invention in this patentdocument are by way of illustration only and should not be construed inany way to limit the scope of the invention. Those skilled in the artwill understand that the principles of the present invention may beimplemented in any suitable manner and in any type of suitably arrangeddevice or system.

FIGS. 1A and 1B illustrate example systems that use multilevel coding(MLC) with binary alphabet polar codes in accordance with thisdisclosure. The systems shown in FIGS. 1A and 1B are for illustrationonly. The techniques described in this patent document could be used tosupport wireless or other communications that use MLC with higher-ordermodulation (such as M-ary quadrature amplitude modulation) and binaryalphabet polar codes between or with any suitable devices or systems.

As shown in FIG. 1A, a system 100 includes base stations 102-104. Eachbase station 102-104 shown in FIG. 1A generally denotes aninfrastructure component that provides wireless service to one or morewireless devices. Other types of infrastructure components that areencompassed by the phrase “base station” include enhanced Node Bs (eNBs)and access points. Similarly, a base station can communicate with anynumber of wireless devices, each of which generally denotes a devicethat receives wireless service from at least one infrastructurecomponent. Other types of devices that are encompassed by the phrase“wireless device” include mobile stations, user equipment, wirelessterminals, or other fixed or mobile devices. The base stations 102-104could communicate using any suitable protocol(s) supporting the use ofpolar codes.

As shown in FIG. 1A, each base station 102-104 includes at least oneprocessing device 106 and at least one memory 108. The at least oneprocessing device 106 executes logic in order to control the overalloperation of the base station 102-104 and to support communications withwireless devices. The exact operations of the processing device(s) 106can vary depending on, for example, the protocol(s) supported by thebase stations 102-104. Each processing device 106 includes any suitableprocessing or control device, such as a microprocessor, microcontroller,digital signal processor (DSP), field programmable gate array (FPGA),application specific integrated circuit (ASIC), or discrete logicdevices. The at least one memory 108 stores instructions and data used,generated, or collected by the base station 102-104. Each memory 108includes any suitable volatile or non-volatile storage and retrievaldevice, such as a random access memory (RAM) or a Flash or otherread-only memory (ROM).

Each base station 102-104 also includes multiple communicationtransceivers 110 a-110 n, each of which (if implemented to supportwireless communications) may be coupled to at least one of multipleantennas 112 a-112 n. Each communication transceiver 110 a-110 nsupports communications with other devices, such as portable or otherwireless devices. Each communication transceiver 110 a-110 n includesany suitable structure supporting communications over one or more wiredor wireless communication channels. For example, each communicationtransceiver 110 a-110 n can include transmit circuitry that facilitatesthe transmission of signals from the communication transceiver andreceive circuitry that facilitates the processing of signals received bythe communication transceiver. Each antenna 112 a-112 n represents anysuitable structure for transmitting and receiving communication signals,such as a radio frequency (RF) antenna. Note that while eachcommunication transceiver 110 a-110 n is shown here as having its ownantenna 112 a-112 n, multiple communication transceivers could share oneor more common antennas, or each communication transceiver could havemultiple antennas.

A backhaul interface 114 supports communications between the basestations 102-104 themselves or between a base station 102-104 and othercomponent(s) via a backhaul network. For example, the backhaul interface114 could allow the base stations 102-104 to communicate with oneanother. The backhaul interface 114 includes any suitable structure forfacilitating communications over a backhaul network, such as a microwavecommunications unit or an optical fiber interface.

FIG. 1B illustrates an example system having a point-to-pointtransmitter 150 and receiver 160. The transmitter 150 and the receiver160 could be used in the base stations 102-104 of FIG. 1A (such as in orwith the processors 106, communication transceivers 110 a-110 n, orbackhaul interfaces 114) or in any other suitable devices. As shown inFIG. 1B, the transmitter 150 includes a forward error correction (FEC)encoder 152, a symbol mapper 154, an analog front end 156, and at leastone antenna 158. The FEC encoder 152 receives information bits andgenerates encoded bits, such as by using MLC with binary alphabet polarcodes as described below. The symbol mapper 154 maps the encoded bitsinto transmit symbols, which the analog front end 156 transmitswirelessly via the antenna(s) 158.

The receiver 160 includes at least one antenna 162, an analog front end164, an equalizer 166, a symbol demapper or soft slicer 168, and an FECdecoder 170. The analog front end 164 receives symbols transmitted fromthe transmitter 150 via the antenna(s) 162 and provides the receivedsymbols to the equalizer 166. The equalizer 166 pre-processes thereceived symbols, and the equalized symbols are used to generatelog-likelihood ratio (LLR) values by the symbol demapper or soft slicer168. The LLR values are used by the FEC decoder 170 to generate decodedinformation bits. Ideally, the decoded information bits from the FECdecoder 170 match the information bits provided to the FEC encoder 152.

Communications from a transmitter to a receiver can occur using variousmodulation techniques, such as binary modulation or M-QAM. Higher-ordermodulation techniques are often desirable because they allow systems toachieve very high levels of spectral efficiency. However, as notedabove, standard techniques for polar code design are not generallyapplicable to systems using higher-order modulation techniques.

This disclosure describes techniques that support the use of MLC withbinary alphabet polar codes, including for use in systems that usehigher-order modulation techniques such as M-ary QAM. The design of aspecific polar code is equivalent to determining a set of “frozen bits”in an input vector. A “frozen bit” denotes a bit whose value is fixed inboth the input vector to a polar encoder and in an output vector from apolar decoder. The remaining bits in the input and output vectors arecalled “information bits” and can be used to transport data over awireless or other channel.

Some conventional approaches use a set of information bits and frozenbits obtained from Monte Carlo simulations that are run for binary inputpolar codes or that are optimized for cases when a binary erasurechannel (BEC) is used. Early work in polar code design focused on binarychannels where the outputs of an encoder are subject to the same binaryinput channel, meaning bits at the output of the encoder experience thesame channel before a decoder. Extending polar codes to higher-orderconstellations may be necessary for deployment in practicalcommunication systems, which usually use higher-order modulation forbetter spectral efficiency. Further, binary codes are usually used inpractice, and the mapping to higher-order constellations is typicallydone after the encoder (rather than using non-binary codes). Althoughthe capacity-achieving property of polar codes has been extended toM-ary input alphabets, the combination of (i) binary polar codes andhigher-order modulations and (ii) specific designs for QAM modulationsin the presence of additive white Gaussian noise (AWGN) has not beenfully investigated. The main difficulty in this case is that the outputbits from a binary polar encoder experience different effective channelsbecause of their mapping to higher-order symbols. Therefore, standardtechniques for polar code design in general are not applicable.

As described in more detail below, this patent document enables the useof binary polar codes with higher-order modulations. A multilevel codingapproach uses one or more binary alphabet polar codes (each possiblywith a different code parameter or configuration such as code rate orfrozen bit locations) with one or more wireless or other communicationchannels, such as one or more AWGN channels. For example, with a specialsymbol mapping rule (such as those shown in FIGS. 2, 4, and 6 describedbelow), one or more channels can be partitioned into m bit levels (alsocalled “bit channels”), which are typically identified based on the bitposition of the symbol word (symbol mapper input). Different bit levelshave different signal-to-noise ratio (SNR) properties. The bits in eachbit level experience the same communication channel with the sameeffective SNR, so the code design problem is reduced to the design ofone or more binary codes with suitable code rate(s). The rate of a polarcode for a particular bit channel can be selected using knowledge of theEuclidian distance for a bit level in a corresponding multilevel pulseamplitude modulation (M-PAM) scheme, assuming bits in lower bit levelsare decoded correctly and known. Thus, the overall error rate (such asthe bit error rate or “BER”) for a polar code is substantiallycharacterized or determined by the required rate of the code at a givenchannel SNR.

Moreover, polarization parameters obtained for true equivalent bitchannels can be used when selecting the rates for the polar codes. Manyconventional approaches that use MLC with polar codes use polarizationparameters optimized with a binary input alphabet for a simple channel(such as a binary erasure channel). In accordance with this disclosure,for higher-order modulation in an AWGN channel, different polarizationparameters can be used for different channels that are “seen” by thedifferent bit levels.

Once the rate of each polar code is identified, data is transmitted overat least one wireless or other channel as symbols created using thepolar code(s). At a receiver, a decoder can be applied at each bitlevel, such as by using successive cancellation (SC) decoders startingwith the least significant bit (LSB).

Various benefits can be obtained using this approach, although thespecific benefits depend on the particular implementation. For example,binary alphabet polar codes can allow for the use of low-complexityencoding and decoding techniques, which can substantially reduce thecomplexity of the overall multilevel coding technique. Also, the use ofpolar codes can provide large performance improvements when used withmultilevel coding techniques compared to approaches that use a singlelevel of coding. When the code rates for all bit channels are optimized,MLC provides better performance than a single binary input alphabetmapped to a higher-order modulation. In addition, it has been shown thatpolar codes do not have an error floor, which means they can be used invarious applications that require higher-order modulations and very lowerror performance. Note that these benefits are unique because they canbe obtained using one or more AWGN channels, while many conventionalapproaches that use MLC with polar codes derive results for binaryerasure channels only.

Additional details regarding the use of MLC with binary alphabet polarcodes are provided below. In the discussion below, two general schemesare described as being used to support MLC with binary alphabet polarcodes. In a first MLC scheme (such as is shown in FIGS. 2 through 5),each binary polar code corresponds to one bit in a bit-to-symbol mappingfor each of the in-phase (I) and quadrature (Q) directions. In a secondMLC scheme (such as is shown in FIGS. 6 and 7), a binary polar codecorresponds to more than one bit in a bit-to-symbol mapping for each ofthe I and Q directions.

Although FIGS. 1A and 1B illustrate examples of systems that use MLCwith binary alphabet polar codes, various changes may be made to FIGS.1A and 1B. For example, MLC with binary alphabet polar codes could beused in any other suitable system. Moreover, infrastructure components,networks, transmitters, and receivers come in a wide variety of designsand configurations, and FIGS. 1A and 1B do not limit the scope of thisdisclosure to any particular infrastructure component, network,transmitter, or receiver.

FIGS. 2 through 7 illustrate example encoding and decoding schemessupporting MLC with binary alphabet polar codes in accordance with thisdisclosure. The encoding and decoding schemes could, for example, beused by the base stations 102-104 or by the transmitter 150 and receiver160 to communicate over one or more wireless or other communicationchannels. However, the encoding and decoding schemes could be used withany other suitable devices and in any other suitable systems.

FIGS. 2 and 3 illustrate a first example implementation of a codingscheme in which each of multiple binary polar codes corresponds to onebit in each of the I and Q directions. FIG. 2 illustrates an examplesymbol mapping 200, which can be used with 16-QAM modulation. Each ofthe I and Q directions independently uses the same mapping as 4-PAM,where (b₁ b₀) denotes the bit mapping to a symbol, with b₁ and b₀representing the most significant bit (MSB) and the least significantbit (LSB), respectively. In FIG. 2, d_(b0) represents the minimumEuclidian distance for b₀, and d_(b1) represents the minimum Euclidiandistance for b₁ (assuming that the bit value for b₀ is correctly decodedand known at the receiver before making a decision for b₁). In otherwords, during decoding, the bit value for b₀ is determined, and thedecoded bit value for b₀ is used to decode the bit value for b₁.

An encoding and decoding system 300 that supports this symbol mapping200 is shown in FIG. 3. As shown in FIG. 3, input vectors u₁ ^(N) 302a-302 b (each containing N bits) are respectively provided to polarencoders 304 a-304 b. The polar encoders 304 a-304 b use the inputvectors 302 a-302 b and different polar codes to generate codewords x₁^(N) 306 a-306 b, respectively, which represent encoded vectors (eachcontaining N bits). As described in more detail below, each polar codeincludes a set of frozen bits with the remaining bits being informationbits. The code rate R₁ for the first polar code used by the polarencoder 304 a is expressed as K₁/N, and code rate R₂ for the secondpolar code used by the polar encoder 304 b is expressed as K₂/N (whereK₁ and K₂ denote the number of information bits for the polar codes).The overall code rate R can be calculated as (K₁+K₂)/2N=(R₁+R₂)/2. At agiven channel condition (such as a given SNR), the K₁ or K₂ value cannotbe increased, which acts as a limit to the code rate for the givenchannel condition. In the discussion below, N_(s) denotes the number ofQAM symbols that carry 2N encoded bits and can equal N/2. The codewords306 a-306 b are provided to a symbol mapper 308, which generates aseries of transmit symbols 310. Each polar encoder 304 a-304 b includesany suitable structure for generating codewords using a binary alphabetpolar code. The symbol mapper 308 includes any suitable structure formapping codewords to symbols.

The symbols 310 are transmitted over at least one communication channel312, which suffers from noise. Each channel 312 could represent anysuitable communication channel, such as a wireless or fiber opticchannel. In some embodiments, each channel 312 could be estimated as anAWGN channel.

The symbols 310, as transported over the channel(s) 312 and affected bynoise, are received as symbols y₁ ^(Ns) 314. LLR calculators 316 a-316 bcompute LLR values for different portions of the received symbols 314 togenerate blocks 318 a-318 b of LLR values. Polar decoders 320 a-320 bdecode the received symbols 314 using the LLR blocks 318 a-318 b and thepolar codes to generate output vectors û₁ ^(N) 322 a-322 b. Ideally, theoutput vectors 322 a-322 b match the input vectors 302 a-302 b. Theoutput vectors 322 a-322 b could be provided to any suitabledestination(s), such as to a common destination or to multipledestinations via de-multiplexing. Each of the polar decoders 320 a-320 bcan support successive cancellation and use feedback in the form of atleast some of the bits (such as the least significant bits) from one ormore prior output vectors 322 a-322 b when decoding one or more currentLLR blocks 318 a-318 b. Each LLR calculator 316 a-316 b includes anysuitable structure for computing LLR values. Each polar decoder 320a-320 b includes any suitable structure for generating decoded bitsusing LLR values and a polar code.

There are various “frozen” bits 324 a-324 b contained in the inputvectors 302 a-302 b and corresponding “frozen” bits 326 a-326 bcontained in the output vectors 322 a-322 b. As noted above, each frozenbit 324 a-324 b, 326 a-326 b denotes a bit whose value is fixed (eitherto a “1” or a “0”) in both the input vector to a polar encoder and inthe output vector from a polar decoder. The remaining bits in the inputvectors 302 a-302 b and the output vectors 322 a-322 b representinformation bits and are used to transport data.

To support multilevel coding, the LLR calculators 316 a-316 b operatesequentially on the received symbols 314, meaning the LLR calculator 316b (on a second bit level) generates LLR values using decoded outputsfrom the polar decoder 320 a (on a first bit level). For example, inresponse to receiving a set of symbols 314, the LLR calculator 316 a canoutput channel LLR blocks 318 a (such as column vectors of length N) tothe polar decoder 320 a. The LLR calculator 316 b uses a new outputvector 322 a from the polar decoder 320 a to generate LLR blocks 318 bat a second bit level.

In the example shown in FIG. 3, the MLC scheme for 16-QAM employs twobinary polar codes. The polar encoder 304 a and the polar decoder 320 asupport a first polar code with a binary alphabet for the first bitlevel, with one bit (b_(0I)) in the I direction and one bit (b_(0Q)) inthe Q direction. The polar encoder 304 b and the polar decoder 320 bsupport a second polar code with a binary alphabet for the second bitlevel, with one bit (b_(1I)) in the I direction and one bit (b_(1Q)) inthe Q direction. Here, when decoding an i^(th) information bit, anylower decoded bits are used, meaning the second information bit isdecoded using the decoded value of the first information bit, the thirdinformation bit is decoded using the decoded values of the first andsecond information bits, and this process can be repeated until all bitsare decoded. Note, however, that other forms of decoding could be used,such as list decoding or belief-propagation decoding. Moreover, notethat this approach can be easily extended to support higher-ordermodulation techniques. An example of this is shown in FIGS. 4 and 5,where this approach has been extended to support 64-QAM modulation.

FIGS. 4 and 5 illustrate a second example implementation of a codingscheme with three bit levels, where different bit levels may beassociated with different polar coding schemes. FIG. 4 illustrates anexample symbol mapping 400, which can be used with 64-QAM modulation.Each of the I and Q directions independently uses the same mapping as8-PAM, where (b₂ b₁ b₀) denotes the bit mapping to a symbol, with b₂ andb₀ representing the MSB and LSB, respectively. In FIG. 4, d_(b0)represents the minimum Euclidian distance for b₀, d_(b1) represents theminimum Euclidian distance for b₁ (assuming that the bit value for b₀ iscorrectly decoded and known at the receiver before making a decision forb₁), and d_(b2) represents the minimum Euclidian distance for b₂(assuming that the bit values for b₀ and b₁ are correctly decoded andknown at the receiver before making a decision for b₂). In other words,during decoding, the bit value for b₀ is determined, the decoded bitvalue for b₀ is used to decode the bit value for b₁, and the decoded bitvalues for b₀ and b₁ are used to decode the bit value for b₂.

An encoding and decoding system 500 that supports this symbol mapping400 is shown in FIG. 5. For higher-order modulations such as 64-QAM, oneapproach is to encode lower bit levels while leaving the MSB uncoded. Anexample of this is shown in FIG. 5, where input vectors u₁ ^(N) 502a-502 b (each containing N bits) are respectively provided to polarencoders 504 a-504 b, while input vectors u₁ ^(N) 502 c (each containingN bits) could remain uncoded. Alternatively, the input vectors 502 ccould be encoded using an encoder 504 c. The encoder 504 c could supportany suitable code, such as a Bose, Chaudhuri and Hocquenghem (BCH) codeto provide a certain level of error protection or a different polarcode.

The polar encoders 504 a-504 b use the input vectors 502 a-502 b anddifferent polar codes to generate codewords x₁ ^(N) 506 a-506 b,respectively. Codewords x₁ ^(N) 506 c contain either the uncoded inputvectors 502 c or the outputs of the encoder 504 c. The codewords 506a-506 c are provided to a symbol mapper 508, which generates a series oftransmit symbols 510. The symbols 510 are transmitted over at least onecommunication channel 512, which suffers from noise.

Received symbols y₁ ^(Ns) 514 are obtained over the channel(s) 512, andLLR calculators 516 a-516 c use the received symbols 514 to generate LLRblocks 518 a-518 c. Polar decoders 520 a-520 b decode portions of thereceived symbols 514 using the LLR blocks 518 a-518 b and the polarcodes to generate output vectors 522 a-522 b. A third decoder 520 cdecodes other portions of the received symbols 514 using the LLR blocks518 c to generate output vectors 522 c. The decoder 520 c couldrepresent any suitable decoder, such as a maximum likelihood (ML) orother hard decoder (when the input vectors 502 c are uncoded) or a BCH,polar, or other decoder (when the input vectors 502 c are coded).

To support multilevel coding, the LLR calculators 516 a-516 c againoperate sequentially on the received symbols 514. For example, inresponse to receiving a set of symbols 514, the LLR calculator 516 a canoutput channel LLR blocks 518 a (such as column vectors of length N) tothe polar decoder 520 a. The LLR calculator 516 b uses a new outputvector 522 a from the polar decoder 520 a to output channel LLR blocks518 b to the polar decoder 520 b. The LLR calculator 516 c uses the newoutput vectors 522 a-522 b from the polar decoders 520 a-520 b to outputchannel LLR blocks 518 c to the decoder 520 c. Note again, however, thatother forms of decoding could be used, such as list decoding orbelief-propagation decoding.

FIGS. 6 and 7 illustrate an example implementation of a coding scheme inwhich one binary polar code may correspond to multiple bits for each ofthe I and Q directions. FIG. 6 illustrates an example symbol mapping600, which can be used with 64-QAM modulation. Each of the I and Qdirections independently uses the same Gray mapping, where (b₂ b₁ b₀)denotes the bit mapping to a symbol, with b₂ and b₀ representing the MSBand LSB, respectively. Note that the Gray mapping is applied only to thetwo least significant bits (b₁ and b₀) in this example, not across alleight constellation points in FIG. 6. This scheme allows large Euclidiandistances for b₂ (MSB) with knowledge of correctly-decoded bits for b₁and b₀. In this embodiment, the two least significant bits are protectedwith the polar code.

An encoding and decoding system 700 that supports this symbol mapping600 is shown in FIG. 7. The encoding and decoding system 700 representsone example of a practical scheme for applying polar codes to MLC withhigher-order modulation and the symbol mapping of FIG. 6. As shown inFIG. 7, input vectors 702 a-702 b are received, where the input vectors702 a contain twice the number of bits (2N) as the input vectors 702 b(N). In the input vectors 702 a, K₁ would equal 2N minus the number offrozen bits. The input vectors 702 a are provided to a polar encoder 704a, which uses the input vectors 702 a and a polar code to generatecodewords 706 a that are mapped to the two LSBs on each of the I and Qdirections (b_(0I), b_(0Q), b_(1I), and b_(1Q)).

The input vectors 702 b may remain uncoded or be subjected to codingusing an encoder 704 b, such as a BCH encoder or another polar encoderwith a high code rate. The uncoded or encoded input vectors 702 brepresent codewords 706 b. The codewords 706 a-706 b are provided to asymbol mapper 708, which generates a series of transmit symbols 710. Thesymbols 710 are transmitted over at least one communication channel 712,which suffers from noise. Received symbols 714 are used by LLRcalculators 716 a-716 b to generate LLR blocks 718 a (for b_(0I),b_(0Q), b_(1I), and b_(1Q)) and LLR blocks 718 b (for b_(2I) andb_(2Q)). A polar decoder 720 a decodes the LLR blocks 718 a using thepolar code to generate output vectors 722 a, while a decoder 720 b (suchas an ML, BCH, or polar decoder) decodes the LLR blocks 718 b intooutput vectors 722 b.

In the approach shown in FIG. 7, the symbols 710 can be grouped intoblocks of size N_(s). For 64-QAM, each symbol uses three bits in the Idirection and three bits in the Q direction. Thus, input vectors 702 aof 2N bits each are encoded using the polar encoder 704 a with the polarcode, and the resulting blocks of 2N bits are multiplexed to the I and Qdimensions for the 64-QAM symbols. The third bit level positions of theI and Q dimensions for the 64-QAM symbols come from the input vectors702 b or the outputs of the encoder 704 b, which have a length of sizeN. Since a Gray coding is used, the LLR values are computed according tothe different mapping. Two or more bits can be decoded together at bitlevels where Gray mapping is used.

Note that FIG. 7 shows the polar encoder 704 a handling the b₀ and b₁bits and the encoder 704 b optionally handling the b₂ bits. Similarly,FIG. 7 shows the polar decoder 720 a handling the b₀ and b₁ bits and thedecoder 720 b handling the b₂ bits. These bit assignments are forillustration only, and other embodiments could be used. For instance,the polar encoder 704 a could handle the b₀ bits, while the encoder 704b handles the b₁ and b₂ bits. In those embodiments, the polar decoder720 a would handle the b₀ bits, and the decoder 720 b would handle theb₁ and b₂ bits.

In general, with respect to any of the embodiments shown in FIGS. 2through 7, a polar code can be characterized by a set of parameters suchas (N, K, A, u_(Ac)). Here, N denotes the output block size in bits, Kdenotes the input information size in bits, A denotes the set of indicesof the information bits (which is complemented by Ac that contains theindices of the frozen bits), and u_(Ac) denotes the values of the frozenbits. A standard generator matrix G_(N) can be used for the encoding,such as:

G _(N) =B _(N) G ₂ ^(⊗n)  (1)

where B_(N) is a bit reversal-based permutation matrix, and G₂ ^(⊗n)denotes the Kronecker product of

$G_{2} = {\begin{bmatrix}1 & 0 \\1 & 1\end{bmatrix}.}$

The polar code can have a length of N=2^(n), with K denoting thedimension of the set of information bits |A|. With the generator matrixdefined in Equation (1), the encoding can be described by:

x ₁ ^(N) =u ₁ ^(N) G _(N)  (2)

where u₁ ^(N) is a row vector of length N of uncoded bits, and x₁ ^(N)denotes a codeword (encoded bits) of length N. The code rate can begiven by the ratio of K/N, while the set of frozen bits and their valueare denoted by the vector u_(Ac) of length N−K.

There are various challenges encountered in the construction of polarcodes, such as when higher-order modulation symbols are sent over anAWGN channel. When M-QAM symbols are used, this can be described as:

y _(k) =s _(k) +n _(k) ,k=1,2, . . . ,N _(s)  (3)

where s_(k) denotes the k^(th) M-QAM symbol sent over the physicalchannel, n_(k) denotes the complex AWGN CN(0, σ²) with zero mean and σ²variance, and N_(s) denotes the number of symbols to transmit a codeblock.

By using a specially-designed symbol mapping rule (such as one shown inFIG. 2, 4, or 6), due to different Euclidean distances between symbolswith 0 or 1 on bit levels from the bit-to-symbol mapping, differentpolar codes have different properties. For example, the second bit levelis better protected than the first bit level, so the effective SNR onthe first bit level is lower than the SNR on the second bit level. As aresult, the code rate R₁=K₁/N (where K₁ is the number of informationbits for the polar code on the first bit level and N is the code blocklength) is desired to be smaller than the code rate R₂=K₂/N of the polarcode applied to the second bit level (where K₂ is the number ofinformation bits for the polar code on the second bit level). Onechallenge in designing such polar codes involves choosing theappropriate code rates such that the error rates of the polar codes atdifferent bit levels are similar or equal (such as within a thresholdamount of one another).

The following represents an example technique for optimizing the coderates for the polar codes at different bit levels, although othertechniques could be used. In one technique, rather than using thetypical construction process of running Monte Carlo simulations toobtain polarization parameters for each SNR evaluation, one ordered setof equivalent channel indexes W_(N) ^((i))(u_(i)|y₁ ^(N),û₁ ^(i−1)) canbe used, where the ordered set is obtained from the SNR value affectingeach bit level. The W_(N) ^((i))(u_(i)|y₁ ^(N),û₁ ^(i−1)) notation usedfor the equivalent channel indexes above is described in Arikan,“Channel Polarization: A Method for Constructing Capacity-AchievingCodes for Symmetric Binary-Input Memoryless Channels,” IEEE Transactionson Information Theory, vol. 55, pp. 3051-3073, July 2009. For example,this can be done by analyzing the tail threshold value of probabilitydensity functions of vectors containing modified log likelihood ratiosmLLR(W_(N) ^((i))), which can be defined as:

$\begin{matrix}{{{mLLR}( W_{N}^{(i)} )} = \{ \begin{matrix}{{+ {{{LLR}( W_{N}^{(i)} )}}},{{if}\mspace{14mu} {correct}\mspace{14mu} {decision}\mspace{14mu} {was}\mspace{14mu} {made}}} \\{{- {{{LLR}( W_{N}^{(i)} )}}},{{if}\mspace{14mu} {wrong}\mspace{14mu} {decision}\mspace{14mu} {was}\mspace{14mu} {made}}}\end{matrix} } & (4)\end{matrix}$

The log-likelihood ratio is defined as:

$\begin{matrix}{{{LLR}( W_{N}^{(i)} )} = {\log ( \frac{W_{N}^{(i)}( {Y_{1}^{N},{ U_{1}^{i - 1} \middle| U_{i}  = 0}} )}{W_{N}^{(i)}( {Y_{1}^{N},{ U_{1}^{i - 1} \middle| U_{i}  = 0}} )} )}} & (5)\end{matrix}$

For a given channel SNR, a BER estimation can be determined using themLLR values as follows. First, from a large set of mLLR data for binarymodulation channels (such as data obtained from simulation), calculate athreshold r^((i))(β) for a target probability of error β. For instance,denoting the distribution of mLLR data as f^((i))(x), the thresholdτ^((i))(β) can be calculated as:

∫_(∞) ^(τ) ^((i)) ^((β)) f ^((i))(x)dx=β  (6)

In other embodiments, the mean and variance of the mLLR values can becalculated, and a Gaussian probability density function (PDF) can beconstructed with the same mean and variance. This can be done using onlysecond-order statistic information (mean and variance). Second, theeffective channel is selected that has τ^((i))(β)<0, and its code ratecan be expressed as:

$\begin{matrix}{R = {1 - {\frac{1}{N}{\sum\limits_{i = 1}^{N}{I\{ {{\tau^{(i)}(\beta)} > 0} \}}}}}} & (7)\end{matrix}$

where the indicator function I{τ^((i))(β)>0} equals one whenτ^((i))(β)>0 and zero otherwise. This approach can be used to identifythe optimal code rate for MLC by selecting the individual bit levels'code rates so that the error rates of the bit levels are approximatelyequal. An example of this can be expressed as:

P _(b) ⁽¹⁾ ≈P _(b) ⁽²⁾ ≈ . . . ≈P _(b) ^((q))≈constant target error rate(β_(target))  (8)

where P_(b) ⁽¹⁾ is the bit error rate for the i^(th) level in theq-level MLC.

In particular embodiments, the following procedure can be used to selectthe polar codes for use in an MLC scheme. In the following, a ratedesign rule is based on keeping substantially the same frame error rate(FER) for each bit level. The set of frozen bits (equivalent to choosingthe coding rate R) is selected for each polar code so that substantiallythe same FER is achieved at the effective SNR experienced at each bitlevel, which can be determined by the corresponding Euclidean distanceof neighboring constellation points in the symbol mapping that is beingused.

A receiver (such as the receiver 160 in FIG. 1B) can use a multistagedecoding strategy, which could be viewed as a higher-level extension ofthe successive cancellation algorithm where there is progressivedecoding of bit levels rather than individual bits. At the decoder, thebits at the first bit level (in both I and Q directions) are decodedfirst. These decoded bits are used to remove constellation points thathave different bit assignments before computing the input LLRs for thebits of the second bit level, and the same process can be repeated withthe higher bit levels. This increases the effective SNR at each bitlevel. The design of MLC polar coding in this approach involves two mainoperations, namely (i) identifying the code rate at each bit level and(ii) designing polar codes with different rates for a given operatingSNR point (which is equivalent to choosing the set of frozen bits).

Consider an MLC configuration with a 2^(2q)-ary digital modulationscheme. Let {K_(j):j=0, q−1} denote the set of information bits at eachbit level, which gives the set of code rates {R_(J):j=0, . . . , q−1} ofthe binary input alphabet polar codes of length N used at each bitlevel. Note that coding across the I and Q axes is symmetric. Let p_(F)^((j))(γ) denote the probability of a frame error at the j^(th) bitlevel (where j=0 for the LSB) at a channel SNR of γ dB. If aspecially-designed symbol mapping rule (such as in FIG. 2, 4, or 6) isused, the minimum distance between adjacent constellation points isdoubled with increased bit levels (assuming all bits at the previous bitlevel are decoded correctly). For the embodiments in FIGS. 2 through 5,this could be viewed as increasing the effective SNR at each bit levelby 6 dB such that the effective SNR γ_(j) at the j^(th) bit level is:

γ_(j) (dB)=γ+6j  (9)

For the embodiments in FIGS. 6 and 7, this can be expressed as:

γ₂ (dB)=γ₁+12  (10)

The set of code rates {R_(j)} can be chosen such that the same FER P_(j)^((j))(γ_(j)) of the corresponding polar code is obtained (or at leastin the same range) among all bit levels, which can be expressed as:

P _(f) ^((j))(γ_(j))≈ P _(F) ,j=0, . . . ,q−1  (11)

where q is the number of bit levels per I/Q direction and P _(F) is thetarget FER that is minimized for the operating SNR. As an example, for64-QAM, 8-PAM is used for both the I and Q components of the signal, andq=3. For the first bit level, the binary mapping of 8-PAM is defined byb_(0I) and b_(0Q) for the I and Q components, respectively. The ratedesign rule defined above states that, for a specific FER of the overallsystem, imposing substantially the same P_(f) ^((j)) for each bitlevel's polar code induces different rates R_(j) because of thedifferent SNR γ_(j) experienced by each bit level j. The overall codedesign problem is therefore reduced to the design of multiple polarcodes at different SNRs, and the procedure described below can be usedto design the polar codes.

A successive cancellation decoder can be used to decode the polar codeat each bit level. Therefore, using the procedure from above, it isstraightforward to show that the overall frame error rate P_(F)(γ) forthe MLC configuration is:

$\begin{matrix}{{P_{F}(\gamma)} \approx {\sum\limits_{j = 0}^{q - 1}{P_{f}^{(j)}( {\gamma + {6j}} )}}} & (12)\end{matrix}$

Solving Equation (12) for the different code rates uses an explicitrelation between the code rate and the frame error rate at all SNRvalues, which may not be available in general. Therefore, a pragmaticapproach can be used that exploits the fact that the performance is ingeneral characterized by the weakest P_(f) ^((j)). Therefore, tomaximize the MLC efficiency for a given target for P_(F)(γ), the coderates for the various bit levels can be selected such that:

P _(f) ⁽⁰⁾(γ)≈P _(f) ⁽¹⁾(γ+6)≈P _(f) ⁽²⁾(γ+12)≈ . . .   (13)

In this case, the target frame error rate for each bit level isP_(F)(γ)/q, and the code rates are chosen according to the effective SNRfor each bit level.

By identifying the operating SNR and FER of each bit level j, the rateR_(j) and the corresponding set of unfrozen bits can be determined asfollows. With this design, the low complexity of the encoder and decoderof binary input alphabet polar codes is maintained while increasingspectral efficiency through sending M-QAM symbols over a channel. Apolar code can be designed for a given operating SNR and FER point usingthe mLLR values from Equation (4). Using, for example, Monte Carlosimulations, the first two moments of the mLLR values (the mean m_(i)and the standard deviation σ_(i)) can be determined. An SC decoder witha random set of information bits can be used for which, at eachequivalent bit channel, all previously decoded bits are assumed to beknown. Note that the use of an SC decoder is optional and that otherdecoder realizations could be used, such as a list decoder or abelief-propagation decoder. Furthermore, the assumption that a channelis an AWGN channel is for illustration only, and other channel typescould be used. If the PDF of the mLLR values is known, the probabilityof error of the equivalent bit channels can be directly evaluated. Forexample, the Gaussian approximation of a PDF can be calculated using thefirst two moments. This approximation is adequately accurate forhighly-positive polarized equivalent bit channels. With a Gaussianapproximation, the probability of bit error at the i^(th) equivalent bitchannel (assuming all other previous bits are decoded properly) can beexpressed as:

$\begin{matrix}{{\beta_{i} = {Q( {m_{i}/\sigma_{i}} )}}{{where}\text{:}}} & (14) \\{{Q(x)}\overset{\Delta}{=}{\frac{1}{\sqrt{2\pi}}{\int_{x}^{\infty}{\exp ( {{- x^{2}}/2} )}}}} & (15)\end{matrix}$

After computing β_(i) for all input bit locations, the indices of theinformation bits A are chosen to correspond to the equivalent bits withthe smallest values of {β_(i)}. The frame error probability P_(F) canthen be calculated as:

$\begin{matrix}{P_{F} = {{1 - {\prod\limits_{i \in A}\; ( {1 - \beta_{i}} )}} \approx {\sum\limits_{i \in A}\beta_{i}}}} & (16)\end{matrix}$

In general, {β_(i)} are functions of the channel SNR γ, but their ordertends to be insensitive to small changes in the SNR as the SNRincreases. This can be important in practical communication systemssince the operating region (the waterfall region of the SNR/BER curve)for a given code rate is typically within 1-2 dB and it may be necessaryto have consistent ordering for that range.

Thus, for example, in a practical communication system that usesadaptive modulation, a single list can be stored that contains theuniversal ordering of the equivalent bit channels within a certain SNRrange. A distinct list can be used for each modulation order. In eachlist, the indices of the bits at the top of the list can be assigned asfrozen bits. The number of frozen bits (which corresponds to the coderate) can be determined by the SNR. This arrangement significantlysimplifies the mechanism of designing the modulation and coding scheme(MCS) of a system. The above design procedure could be combined with anymodulation scheme, where the modulation and the corresponding LLRcalculation of the individual bits within each symbol are consideredpart of the underlying channel. With the above parameterization, thecode rate |A|/N for a frame error rate P _(F) could be computedanalytically. For a given P _(F), the maximum code rate can correspondto a value calculated as follows:

$\begin{matrix}{{\max {A}{such}\mspace{14mu} {that}{\sum\limits_{i \in A}\beta_{i}}} \leq {\overset{\_}{P}}_{F}} & (17)\end{matrix}$

where it is assumed that the indices in A are sorted in ascending orderof β_(i).

One advantage of the proposed solution described above is the use of“true” polarization parameters for choosing the sets of frozen bits andinformation bits. This can be particularly useful, for example, whenmore than one communication channel is used for physical transmission.One limiting constraint of conventional polar codes is that the samephysical channel is to be used for each equivalent bit channel. However,two or more channels can still provide polarization to the equivalentchannels when the initial G₂ matrix is used as the base for thegenerator matrix. This means that the equivalent channels with thehighest polarization levels can be used for transmitting the informationbits, instead of some mismatched sets that are optimized for differentsystem properties.

Note that the symbol mapping rules (such as the ones shown in FIGS. 2,4, and 6) can introduce a desired property, such as increasing minimumEuclidian distance when moving to higher bit-levels. Gray mapping-basedsymbol mappings typical do not give such a desired property.

The components shown in FIGS. 3, 5, and 7 could be implemented in anysuitable manner. For example, the components shown in FIGS. 3, 5, and 7could be implemented using only hardware components or a combination ofhardware and software/firmware instructions. As particular examples, thecomponents shown in FIGS. 3, 5, and 7 could be implemented using one ormore FPGAs, ASICs, or other hardware-only components. The componentsshown in FIGS. 3, 5, and 7 could also be implemented using one or moreprocessing devices that execute software/firmware instructions forperforming the described functions. A combination of approaches couldalso be used, such as when some components are implemented using onlyhardware and other components are implemented using one or moreprocessing devices that execute software/firmware instructions.

Although FIGS. 2 through 7 illustrate examples of encoding and decodingschemes supporting MLC with binary alphabet polar codes, various changesmay be made to FIGS. 2 through 7. For example, this disclosure is notlimited to the use of two polar codes at two bit levels of an MLCscheme. Two or more binary alphabet polar codes can be used at two ormore bit levels of an MLC scheme.

FIGS. 8A and 8B illustrate example methods for using MLC with binaryalphabet polar codes in accordance with this disclosure. In particular,FIG. 8A illustrates an example method 800 that could be used by anencoder supporting MLC, and FIG. 8B illustrates an example method 850that could be used by a decoder supporting MLC. These methods 800 and850 could, for example, be used by the base stations 102-104 or by thetransmitter 150 and receiver 160 to communicate over one or morewireless or other communication channels. However, the methods 800 and850 could be used with any other suitable devices and in any othersuitable systems.

As shown in FIG. 8A, data bits to be transmitted are received at step802. This could include, for example, receiving one or more inputvectors 302 a-302 b, 502 a-502 c, 702 a-702 b containing data bits to betransmitted. One or more polar codes are applied to one or more subsetsof the bits to generate codewords at step 804. This could include, forexample, the polar encoders 304 a-304 b, 504 a-504 b applying differentpolar codes to different input vectors 302 a-302 b, 502 a-502 b. Thiscould also include the polar encoder 704 a applying a polar code to theinput vectors 702 a. Note that some bits, such as most significant bits,may or may not be encoded during the generation of the codewords.

Symbols are generated using the codewords at step 806. This couldinclude, for example, the symbol mapper 308, 508, 708 mapping thecodewords to different QAM symbols based on an appropriate symbolmapping, such as the symbol mapping 200, 400, or 600. The symbols aretransmitted over one or more communication channels at step 808. Thiscould include, for example, transmitting the symbols over one or moreAWGN channels.

As shown in FIG. 8B, symbols transmitted over one or more communicationchannels are received at step 852. This could include, for example,receiving the symbols over one or more AWGN channels. Different sets ofdecoding information are extracted from the received symbols at step854. This could include, for example, the LLR calculators 316 a-316 b,516 a-516 c, 716 a-716 b identifying LLR values using the receivedsymbols. The LLR calculators can operate sequentially such that the LLRcalculator on one bit level generates LLR values using decoded outputsfrom the decoder on at least one earlier bit level.

One or more polar codes are applied to one or more of the different setsof decoding information to generate decoded bits at step 856. This couldinclude, for example, the polar decoders 320 a-320 b, 520 a-520 bapplying different polar codes to different LLR blocks 318 a-318 b, 518a-518 b at different bit levels. This could also include the polardecoder 720 a applying a polar code to the LLR blocks 718 a. In eithercase, the coding rate(s) of the polar code(s) allow(s) the different bitlevels to have substantially similar error rates. Note that some bits(such as most significant bits at an additional bit level) could bedecoded in other ways (such as by using an ML or BCH decoder). Thedecoded data bits at the bit different bit levels are output at step858.

Although FIGS. 8A and 8B illustrate examples of methods for using MLCwith binary alphabet polar codes, various changes may be made to FIGS.8A and 8B. For example, while shown as a series of steps, various stepsin each figure could overlap, occur in parallel, or occur any number oftimes.

FIG. 9 illustrates an example method 900 for designing an MLC codingscheme with binary alphabet polar codes in accordance with thisdisclosure. The method 900 could, for example, be performed offline oneor more times prior to creating an encoding or decoding design.

As shown in FIG. 9, a target error rate is identified at step 902. Thiscould include, for example, identifying a desired target FER forcommunications. The desired target FER could be based on the specificapplication or any other suitable criteria. An effective SNR at each ofmultiple bit levels associated with a particular symbol mapping rule isidentified at step 904. As noted above, the symbol mapping rule may bedesigned to have a property that provides increased Euclidian distanceas one goes to higher bit levels in MLC with knowledge ofcorrectly-decoded bits for earlier bit level(s). For example, as notedabove, the symbol mappings could provide a known increase to theeffective SNR at each bit level, such as by 6 dB or 12 dB.

The code rate at each bit level is determined using the target errorrate and the SNR at that bit level at step 906, and the polar codes aredesigned with the identified code rates at step 908. This could include,for example, selecting the code rates as described above so that theoverall error rate of each polar code is substantially equivalent. Thiscould also include using the mLLR values from Equation (4), determiningtheir mean m_(i) and standard deviation σ_(i), constructing GaussianPDFs using these values, estimating error rates using the PDFs, andselecting the bit locations with the worst error rates as the frozenbits.

Although FIG. 9 illustrates one example of a method 900 for designing anMLC coding scheme with binary alphabet polar codes, various changes maybe made to FIG. 9. For example, while shown as a series of steps,various steps in FIG. 9 could overlap, occur in parallel, occur in adifferent order, or occur any number of times. Also, note that whilemultiple polar codes are designed in FIG. 9, a single polar code couldbe designed, such as for use in the system 700 of FIG. 7.

FIG. 10 illustrates an example device 1000 supporting the design of anMLC coding scheme with binary alphabet polar codes in accordance withthis disclosure. The device 1000 could, for example, be used toimplement the method 900 shown in FIG. 9 in order to design an MLCcoding scheme with binary alphabet polar codes. Note, however, that themethod 900 could be performed using other devices.

As shown in FIG. 10, the device 1000 includes a bus system 1002. The bussystem 1002 is configured to support communication between at least oneprocessing device 1004, at least one storage device 1006, at least onecommunications unit 1008, and at least one input/output (I/O) unit 1010.

The processing device 1004 is configured to execute instructions thatcan be loaded into a memory 1012. The device 1000 can include anysuitable number(s) and type(s) of processing devices 1004 in anysuitable arrangement. Example processing devices 1004 can includemicroprocessors, microcontrollers, digital signal processors, fieldprogrammable gate arrays, application specific integrated circuits, anddiscrete circuitry. The processing device(s) 1004 can be configured toexecute processes and programs resident in the memory 1012.

The memory 1012 and a persistent storage 1014 can represent anystructure(s) capable of storing and facilitating retrieval ofinformation (such as data, program code, or other suitable informationon a temporary or permanent basis). The memory 1012 can represent arandom access memory or any other suitable volatile or non-volatilestorage device(s). The persistent storage 1014 can contain one or morecomponents or devices supporting longer-term storage of data, such as aread-only memory, hard drive, Flash memory, or optical disc.

The communications unit 1008 is configured to support communicationswith other systems or devices. For example, the communications unit 1008can include a network interface card or a wireless transceiverfacilitating communications over a network. The communications unit 1008can be configured to support communications through any suitablephysical or wireless communication link(s).

The I/O unit 1010 is configured to allow for input and output of data.For example, the I/O unit 1010 can be configured to provide a connectionfor user input through a keyboard, mouse, keypad, touchscreen, or othersuitable input device. The I/O unit 1010 can also be configured to sendoutput to a display, printer, or other suitable output device.

Although FIG. 10 illustrates one example of a device 1000 supporting thedesign of an MLC coding scheme with binary alphabet polar codes, variouschanges may be made to FIG. 10. For example, various components in FIG.10 could be combined, further subdivided, or omitted and additionalcomponents could be added according to particular needs. In general,computing devices can come in a wide variety of configurations, fromservers to desktop or laptop computers to portable devices such asmobile smartphones. FIG. 10 does not limit this disclosure to anyparticular computing device.

In some embodiments, various functions described in this patent documentare implemented or supported by a computer program that is formed fromcomputer readable program code and that is embodied in a computerreadable medium. The phrase “computer readable program code” includesany type of computer code, including source code, object code, andexecutable code. The phrase “computer readable medium” includes any typeof medium capable of being accessed by a computer, such as read onlymemory (ROM), random access memory (RAM), a hard disk drive, a compactdisc (CD), a digital video disc (DVD), or any other type of memory. A“non-transitory” computer readable medium excludes wired, wireless,optical, or other communication links that transport transitoryelectrical or other signals. A non-transitory computer readable mediumincludes media where data can be permanently stored and media where datacan be stored and later overwritten, such as a rewritable optical discor an erasable memory device.

It may be advantageous to set forth definitions of certain words andphrases used throughout this patent document. The terms “application”and “program” refer to one or more computer programs, softwarecomponents, sets of instructions, procedures, functions, objects,classes, instances, related data, or a portion thereof adapted forimplementation in a suitable computer code (including source code,object code, or executable code). The terms “transmit,” “receive,” and“communicate,” as well as derivatives thereof, encompasses both directand indirect communication. The terms “include” and “comprise,” as wellas derivatives thereof, mean inclusion without limitation. The term “or”is inclusive, meaning and/or. The phrase “associated with,” as well asderivatives thereof, may mean to include, be included within,interconnect with, contain, be contained within, connect to or with,couple to or with, be communicable with, cooperate with, interleave,juxtapose, be proximate to, be bound to or with, have, have a propertyof, have a relationship to or with, or the like. The phrase “at leastone of,” when used with a list of items, means that differentcombinations of one or more of the listed items may be used, and onlyone item in the list may be needed. For example, “at least one of: A, B,and C” includes any of the following combinations: A, B, C, A and B, Aand C, B and C, and A and B and C.

While this disclosure has described certain embodiments and generallyassociated methods, alterations and permutations of these embodimentsand methods will be apparent to those skilled in the art. Accordingly,the above description of example embodiments does not define orconstrain this disclosure. Other changes, substitutions, and alterationsare also possible without departing from the spirit and scope of thisdisclosure, as defined by the following claims.

What is claimed is:
 1. A method comprising: applying, by at least oneprocessing device of a wireless device, a first binary alphabet polarcode to a first subset of data bits, to produce first encoded bits, thefirst encoded bits associated with a first bit level of a multilevelcoding scheme; generating, by the at least one processing device, one ormore symbols using the first encoded bits and bits associated with asecond bit level of the multilevel coding scheme; and transmitting, byat least one antenna of the wireless device, the one or more symbols. 2.The method of claim 1, further comprising applying a second binaryalphabet polar code to a second subset of the data bits, to producesecond encoded bits, the second encoded bits associated with the secondbit level of the multilevel coding scheme.
 3. The method of claim 2,wherein the second binary alphabet polar code is associated with asecond coding rate such that the first and second bit levels havesubstantially equal error rates.
 4. The method of claim 1, wherein: thefirst bit level represents one or more least significant bits of the oneor more symbols; and the second bit level represents one or more otherbits of the one or more symbols.
 5. The method of claim 1, wherein thefirst binary alphabet polar code corresponds to one bit in abit-to-symbol mapping for an in-phase (I) direction and one bit in thebit-to-symbol mapping for a quadrature (Q) direction.
 6. The method ofclaim 1, wherein the first binary alphabet polar code corresponds tomultiple bits in a bit-to-symbol mapping for an in-phase (I) directionand multiple bits in the bit-to-symbol mapping for a quadrature (Q)direction.
 7. The method of claim 1, wherein generating the one or moresymbols comprises using a symbol mapping of bits to symbols, the symbolmapping providing an increasing Euclidian distance between symbolshaving a common value at a specific bit position as bit level increases.8. A wireless device, comprising: a polar encoder configured to apply afirst binary alphabet polar code to a first subset of data bits, toproduce first encoded bits, the first encoded bits associated with afirst bit level of a multilevel coding scheme; a symbol mapper coupledto the polar encoder, the symbol mapper configured to map the firstencoded bits and bits associated with a second bit level of themultilevel coding scheme to one or more symbols; and an antenna coupledto the symbol mapper, the antenna configured to transmit the one or moresymbols.
 9. The wireless device of claim 8, further comprising a polarsecond encoder configured to apply a second binary alphabet polar codeto a second subset of the data bits to generate second encoded bits, thesecond encoded bits associated with the second bit level of themultilevel coding scheme.
 10. The wireless device of claim 9, whereinthe second binary alphabet polar code is associated with a second codingrate such that the first and second bit levels have substantially equalerror rates.
 11. The wireless device of claim 8, wherein: the first bitlevel represents one or more least significant bits of the one or moresymbols; and the second bit level represents one or more other bits ofthe one or more symbols.
 12. The wireless device of claim 8, wherein thefirst binary alphabet polar code corresponds to one bit in abit-to-symbol mapping for an in-phase (I) direction and one bit in thebit-to-symbol mapping for a quadrature (Q) direction.
 13. The wirelessdevice of claim 8, wherein the first binary alphabet polar codecorresponds to multiple bits in a bit-to-symbol mapping for an in-phase(I) direction and multiple bits in the bit-to-symbol mapping for aquadrature (Q) direction.
 14. The wireless device of claim 8, whereinthe symbol mapper is further configured to use a symbol mapping of bitsto symbols, the symbol mapping providing an increasing Euclidiandistance between symbols having a common value at a specific bitposition as bit level increases.
 15. A method comprising: receiving, byan antenna of a wireless device, one or more symbols; applying, by atleast one processing device of the wireless device, a first binaryalphabet polar code to first decoding information associated with theone or more symbols, to produce first decoded bits, the first decodinginformation associated with a first bit level of a multilevel codingscheme; and outputting the first decoded bits and bits associated with asecond bit level of the multilevel coding scheme.
 16. The method ofclaim 15, further comprising applying a second binary alphabet polarcode to second decoding information associated with the one or moresymbols to generate second decoded bits, the second decoded bitsassociated with the second bit level.
 17. The method of claim 16,wherein the second binary alphabet polar code is associated with asecond coding rate such that the first and second bit levels havesubstantially equal error rates.
 18. The method of claim 17, furthercomprising calculating log-likelihood ratio (LLR) values associated withthe bit levels using the one or more symbols to generate the first andsecond decoding information, wherein the LLR values associated with thesecond bit level are calculated using the first decoded bits.
 19. Themethod of claim 16, further comprising applying a third decoding schemeto third decoding information associated with the one or more symbols togenerate third decoded bits, the third decoding information associatedwith a third bit level.